Method and apparatus for performing optical imaging using frequency-domain interferometry

ABSTRACT

An apparatus and method are provided. In particular, at least one first electro-magnetic radiation may be provided to a sample and at least one second electro-magnetic radiation can be provided to a non-reflective reference. A frequency of the first and/or second radiations varies over time. An interference is detected between at least one third radiation associated with the first radiation and at least one fourth radiation associated with the second radiation. Alternatively, the first electro-magnetic radiation and/or second electro-magnetic radiation have a spectrum which changes over time. The spectrum may contain multiple frequencies at a particular time. In addition, it is possible to detect the interference signal between the third radiation and the fourth radiation in a first polarization state. Further, it may be preferable to detect a further interference signal between the third and fourth radiations in a second polarization state which is different from the first polarization state. The first and/or second electro-magnetic radiations may have a spectrum whose mean frequency changes substantially continuously over time at a tuning speed that is greater than 100 Tera Hertz per millisecond.

CROSS REFERENCE TO RELATED APPLICATION(S)

The present application is a continuation of U.S. patent applicationSer. No. 13/720,507 filed on Dec. 19, 2012, which is a continuation ofU.S. patent application Ser. No. 13/191,885 filed on Jul. 27, 2011 whichissued as U.S. Pat. No. 8,355,138 on Jan. 15, 2013, which is acontinuation of U.S. patent application Ser. No. 12/795,529 filed onJun. 7, 2010 which issued as U.S. Pat. No. 8,384,909 on Feb. 26, 2013,which is a continuation of U.S. patent application Ser. No. 10/577,562filed Apr. 27, 2006 which issued as U.S. Pat. No. 7,733,497 on June 8,2010, which is a national phase of PCT/US2004/029148 filed on Sep. 8,2004, the entire disclosures of which are incorporated herein byreference. This application also claims priority from U.S. ProvisionalPatent Application Ser. No. 60/514,769 filed on Oct. 27, 2003, theentire disclosure of which is incorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates generally optical imaging, and moreparticularly to method and apparatus for performing optical imagingusing frequency-domain interferometry.

BACKGROUND OF THE INVENTION

As is known in the art, optical interferometric reflectometry is apowerful tool for performing non-invasive, high-resolution (˜10 μm),cross-sectional imaging of a biological or other sample, to visualizemicro-structural optical properties such as reflection, absorption,scattering, attenuation, birefringence, and spectroscopic analysis.There are a number of interferometric imaging techniques that are knownin the art. These techniques in general can be divided into two majorcategories: (i) time-domain technique, and (ii) frequency-domaintechnique.

Low coherence interferometry (“LCI”) is one of the time-domaintechniques. This technique uses a scanning system to vary the referencearm length and acquire the interference signal at a detector. Then, thefringe pattern is demodulated to obtain the coherence envelope of thesource cross correlation function. Optical coherence tomography (“OCT”)is a technique for obtaining two- or three-dimensional images using LCI.OCT is described in U.S. Pat. No. 5,321,501 issued to Swanson et al.Multiple variants of the OCT techniques have been described, but manysuffer from less than optimal signal to noise ratio (“SNR”), resultingin non-optimal resolution, low imaging frame rates, and poor depth ofpenetration. Power usage is a factor in such imaging techniques. Forexample in ophthalmic uses, only a certain number of milliwatts of powerare tolerable before thermal damage can occur. Thus, boosting power isnot feasible to increase SNR in such environments. Nevertheless, itwould be desirable to have an imaging method with superior SNR withoutappreciably increasing power requirements.

Insufficient SNR can also prevent the OCT technique from being used at ahigh frame rate which is important to avoid motion artifacts andovercome the short measurement time window available, for example, forin-vivo vascular imaging. Therefore, a way to improve SNR and imagingspeed (e.g., the frame rate) is desired.

Spectral interferometry, or spectral radar, is one of thefrequency-domain imaging techniques. In spectral radar, the real part ofthe cross spectral density of sample and reference arm light is measuredwith a spectrometer. Depth profile information can be encoded on thecross-spectral density modulation.

The use of spectral radar concepts to increase SNR of LCI and OCT hasbeen described previously. This technique uses a charge coupled device(“CCD”) with a large number of pixels (an order of 1,000) to reach scanranges on the order of a millimeter. The fast readout of the CCD devicemakes high-speed imaging possible.

There are, however, a number of disadvantages associated with using aCCD device. First, CCD devices are relatively expensive compared to asingle-element photo-receiver. Secondly, the previously described methoduses a single CCD to acquire the data. Since the charge storage capacityis limited, it requires a reduction of the reference arm power toapproximately the same level as the sample arm power, giving rise toauto correlation noise on the sample arm light. In addition, since nocarrier is generated, the 1/f noise will dominate the noise in thissystem. Thirdly, even with the short integration times of state of theart CCD technology, phase instabilities in the interferometer reducefringe visibility of the cross spectral density modulation. Thisshortcoming makes the technique vulnerable to motion artifacts.

Coherent frequency-modulated continuous-wave reflectometry (C-FMCW) isanother frequency domain technique known in the art. U.S. Pat. Nos.5,956,355 and 6,160,826 issued to Swanson et al. describes an opticalimaging method and apparatus using this technique. The previouslydescribed imaging method is based on using a continuously-tunedsingle-frequency laser as an optical source. The tuning wavelength rangeis required to be several tens of nanometers to achieve rangingresolution of less than 100 microns. The instantaneous linewidth of thelaser must be less than approximately 0.1 nm to achieve a detectionrange on the order of 1.0 mm. The tuning rate should be greater than 10kHz for high speed (e.g., video-rate) imaging. Although anexternal-cavity semiconductor laser can be configured to achievemode-hop-free single-frequency tuning over several tens of nanometer,the tuning rate has been less than 1 Hz due to stringent requirement onmechanical stability. A way to overcome this speed difficulty ispreferable.

It would, therefore, be desirable to provide a system and method toovercome the source availability and scan speed shortcomings ofconventional LCI and OCT.

SUMMARY OF THE INVENTION

In accordance with exemplary embodiments of the present invention, anexemplary optical frequency domain imaging (“OFDI”) system can include amultiple-frequency-mode (or multiple longitudinal or axial-mode)wavelength-swept laser source optically coupled to an interferometercontaining a sample under study. The system can further include anarrangement which is configured to produce interferometric signals inquadrature between light reflected from a sample and a reference lightand a detector disposed to receive the interferometric signals.

With such exemplary particular arrangement, an OFDI system which canoperate with source powers that are relatively low compared with sourcepowers of conventional systems and/or which operate at acquisition rateswhich are relatively high compared with acquisition rates ofconventional systems may be provided. The use of a swept source resultsin an imaging system having reduced shot noise and other forms of noisewhich allows for much lower source powers, or much higher acquisitionrates than conventional systems. This can lead to an increased detectionsensitivity which results in the ability to provide real time imaging.Such imaging speed can assist practitioners in gastrointestinal,ophthalmic and arterial imaging fields, where motion artifacts are acontinuing problem. By increasing a frame rate while maintaining orimproving the signal to noise ratio such artifacts can be minimized orin some cases eliminated. Exemplary embodiments of the present inventionmay also enable the screening of large areas of tissues with OFDI andallows enables the use of clinically viable screening protocols.

In one exemplary embodiment of the present invention, thewavelength-swept laser can be provided that may use an optical band-passscanning filter in the laser cavity to produce a rapidly-sweptmultiple-frequency-mode output. By using an optical band-pass scanningfilter in the laser cavity, it is not necessary to tune the laser cavitylength to provide synchronous tuning of the laser spectrum. In otherwords, it does not require tuning the longitudinal cavity mode of thelaser at the same rate as the center wavelength of the laser.

In another exemplary embodiment of the present invention, the detectorcan be a dual-balanced receiver disposed to accept interferometricsignals and to suppress the relative intensity noise in theinterferometric signals.

The gain in signal-to-noise ratio (“SNR”) according to an exemplaryembodiment of the present invention is advantageous over time-domainapproaches such as OCT via a performance of the signal processing in theFourier-domain. The SNR enhancement is by a factor of N, the ratio ofthe depth range to the spatial resolution. The enhancement factor N canreach a few hundreds to several thousand. This increase in SNR enablesthe imaging by a factor of N times faster, or alternatively allowsimaging at the same speed with a source that has N times lower power. Asa result, the exemplary embodiment of the present invention overcomestwo important shortcomings of conventional LCI and OCT, e.g., sourceavailability and scan speed. The factor N may reach more than 1,000, andallows construction of OFDI systems that can be more than three ordersof magnitude improved from OCT and LCI technology currently in practice.

The gain in SNR is achieved because, e.g., the shot noise has a whitenoise spectrum. The signal intensity present at the detector atfrequency w (or wavelength λ) contributes only to the signal atfrequency w, but the shot noise is generated at all frequencies. Bynarrowing the optical band width per detector, the shot noisecontribution at each frequency can be reduced, while the signalcomponent remains the same.

Exemplary embodiments according to the present invention improvescurrent data acquisition speeds and availability of sources comparedwith OCT. Shot noise is due to the statistical fluctuations of thecurrent that are due to the quantized or discrete electric charges. Thereduction of shot noise allows for much lower source powers or muchhigher acquisition rates. Limitations in current data acquisition rates(˜4 frames/sec) are imposed by available source power and availabilityof fast mechanisms for scanning delay. An increase in the sensitivity ofthe detection by a factor of 8 would allow real time imaging at a speedof about 30 frames per second. An increase of the sensitivity by afactor of about 1,000-2,000 allows for the use of sources with muchlower powers and higher spectral bandwidths which are readily available,cheaper to produce, and can generate higher resolution OFDI images.

For ophthalmic applications of OFDI, efficient detection preferablyallows for a significant increase of acquisition speed. One limitationin ophthalmic applications is the power that is allowed to enter the eyeaccording to the ANSI standards (approximately 700 microwatts at 830nm). Current data acquisition speed in ophthalmic applications isapproximately 100-500 A-lines per second. The power efficient detectiontechnique of the present invention would allow for A-line acquisitionrates on the order of about 100,000 A-lines per second, or video rateimaging at about 3,000 A-lines per image.

To achieve at least some of the goals of the present invention, anapparatus and method according an exemplary embodiment of the presentinvention are provided. In particular, at least one firstelectro-magnetic radiation may be provided to a sample and at least onesecond electro-magnetic radiation can be provided to a non-reflectivereference. A frequency of the first and/or second radiations varies overtime. An interference is detected between at least one third radiationassociated with the first radiation and at least one fourth radiationassociated with the second radiation. Alternatively, the firstelectro-magnetic radiation and/or second electro-magnetic radiation havea spectrum which changes over time. The spectrum may contain multiplefrequencies at a particular time. In addition, it is possible to detectthe interference signal between the third radiation and the fourthradiation in a first polarization state. Further, it may be preferableto detect a further interference signal between the third and fourthradiations in a second polarization state which is different from thefirst polarization state. The first and/or second electro-magneticradiations may have a spectrum whose mean frequency changessubstantially continuously over time at a tuning speed that is greaterthan 100 Tera Hertz per millisecond.

In one exemplary embodiment of the present invention, the thirdradiation may be a radiation returned from the sample, and the at leastone fourth radiation is a radiation returned from the reference. Thefrequency of the first, second, third and/or fourth radiation may beshifted. An image can be generated based on the detected interference. Aprobe may be used which scans a transverse location of the sample togenerate scanning data, and provides the scanning data to the thirdarrangement so as to generate the image. The scanning data may includethe detected interference obtained at multiple transverse locations onthe sample. At least one photodetector and at least one electricalfilter may be used which follow a photodetector, which is followed by anelectrical filter. The electric filter ma be a bandpass filter having acenter frequency that is approximately the same as a magnitude of thefrequency shift by the frequency shifting arrangement. A transmissionprofile of the electrical filter can vary substantially over itspassband. The probe may include a rotary junction and a fiber-opticcatheter. The catheter can be rotated at a speed higher than 30revolutions per second. At least one polarization modulator may beprovided.

At least one polarization diverse receive and/or a polarization diverseand dual balanced receiver may be used. It is further possible to trackthe phase difference between:

-   -   the first electromagnetic radiation and the second        electromagnetic radiation, and/or    -   the third electromagnetic radiation and the fourth        electromagnetic radiation.

According to still another exemplary embodiment of the presentinvention, the first and second electro-magnetic radiations can beemitted, at least one of which has a spectrum whose mean frequencychanges substantially continuously over time at a tuning speed that isgreater than 100 Tera Hertz per millisecond.

According to still a further exemplary embodiment of the presentinvention, an apparatus is provided. Such apparatus includes at leastone first arrangement providing at least one first electro-magneticradiation to a sample and at least one second electro-magnetic radiationto a reference. The apparatus also includes t least one secondarrangement adapted for shifting the frequency of the firstelectro-magnetic radiation and the second electromagnetic radiation, andan interferometer interfering the first and second electro-magneticradiations to produce an interference signal. Further, the apparatusincludes at least one second arrangement detecting the interferencebetween the first and second electro-magnetic radiations.

Further, according to another exemplary embodiment of the presentinvention, a system, method, software arrangement and storage medium areprovided for determining particular data associated with at least one ofa structure and composition of a tissue. In particular, informationassociated with an interferometric signal is received which is formedfrom at least one first electro-magnetic radiation obtained from asample and at least one second electro-magnetic radiation obtained froma reference. The first and/or second electro-magnetic radiations is/arefrequency-shifted. The information is sampled to generate sampled datain a first format. Further, the sampled data is transformed into theparticular data that is in a second format, the first and second formatbeing different from one another.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention and itsadvantages, reference is now made to the following description, taken inconjunction with the accompanying drawings, in which:

FIG. 1 is a block diagram of a time-domain optical coherence tomography(“OCT”) system;

FIG. 2 is a block diagram of a system which performs frequency-domainimaging using a spectral radar technique;

FIG. 3A is a block diagram of a system which performs frequency-domainimaging using a coherent single-frequency tuning source according to oneexemplary embodiment of the present invention;

FIGS. 3B and 3C are graphs of wavelength versus amplitude which takentogether illustrate the occurrence of a frequency shift generated by thesystem of FIG. 3A;

FIG. 3D is a graph of a beat signal generated by the system of FIG. 3A;

FIG. 4A is a block diagram of a system which performs frequency-domainimaging using a multiple-longitudinal-mode wavelength-swept sourceaccording to another exemplary embodiment of the present invention;

FIGS. 4B and 4C are graphs of a wavelength spectrum which taken togetherillustrate the occurrence of a shift in the spectrum generated by thesystem of FIG. 4A;

FIG. 4D is a graph of a beat signal generated by the system of FIG. 4A;

FIG. 5 is a block diagram of a system which performs frequency-domainimaging using a wavelength-swept source according to another exemplaryembodiment of the present invention;

FIG. 6 is a block diagram of an optical wavelength tunable filterarrangement according to an exemplary embodiment of the presentinvention;

FIG. 7 is a block diagram of a wavelength-swept laser arrangementaccording to an exemplary embodiment of the present invention;

FIG. 8A is an exemplary graph of a laser output spectrum as measured atan output of the wavelength-swept laser arrangement of FIG. 7;

FIG. 8B is an exemplary graph of a laser output as measured at an outputof the wavelength-swept laser of FIG. 7;

FIG. 9A is a block diagram of a wavelength tunable filter arrangementbearing a polygonal mirror according to yet another exemplary embodimentof the present invention;

FIG. 9B is a block diagram of a wavelength tunable filter arrangementhaving reflective disk according to still another exemplary embodimentof the present invention;

FIG. 10A is a block diagram of an optical frequency domain imaging(“OFDI”) system which includes a wavelength-swept laser and apolarization diversity-balanced detection (“PDBD”) circuit according toa further exemplary embodiment of the present invention;

FIG. 10B is a block diagram of an exemplary probe arrangement shown inFIG. 10A;

FIG. 10C is a plurality of graphs illustrating exemplary outputs of acarrier-frequency heterodyne detection using the system of FIG. 10A;

FIG. 11 is an exemplary in-vivo image of a human finger tip obtainedusing exemplary embodiments of the present invention;

FIG. 12 is a block diagram of a phase tracker arrangement according toan exemplary embodiment of the present invention;

FIG. 13 is a block diagram of an exemplary embodiment of the OFDI systemaccording to the present invention having the phase tracker;

FIGS. 14A-14C are flow diagrams which illustrate an exemplary techniquefor a phase tracker operation according to the present invention;

FIG. 15 is a simplified diagram of the OFDI system according to anotherexemplary embodiment of the present invention;

FIGS. 16( a) and 16(b) are graphs of effects of a frequency shiftaccording to the present invention, i.e., depth versus signal frequency;

FIG. 17 is a block diagram of the OFDI system employing twoacousto-optic frequency shifters according to still another exemplaryembodiment of the present invention;

FIGS. 18( a) and 18(c) are graphs of point spread functions measuredwithout a mapping process according to the present invention;

FIGS. 18( b) and 18(d) are graphs of point spread functions measuredwith the mapping process according to the present invention; and

FIGS. 19A and 19B are exemplary illustrations of images and/or graphs ofexperimental results obtained using exemplary embodiments of the presentinvention.

Throughout the drawings, the same reference numerals and characters,unless otherwise stated, are used to denote like features, elements,components, or portions of the illustrated embodiments. Moreover, whilethe present invention will now be described in detail with reference tothe Figures, it is done so in connection with the illustrativeembodiments.

DETAILED DESCRIPTION

FIG. 1 shows an exemplary prior art time domain optical coherencetomography (“OCT”) system 10 which includes a broadband source 12 thatprovides a signal to a first arm 14 a of two-by-two splitter 14. Thesplitter divides the signal provided thereto at port 14 a, and providesa first portion of the signal at a port 14 b coupled to a reference arm16. The splitter 14 also provides a second portion of the signal at aport 14 c coupled to a sample arm 18.

The sample arm 18 terminates at a sample volume 19 and an arrangement 22for providing a lateral scan of the sample volume is disposed in thesample arm 18 prior to the sample volume 19. The reference arm 16terminates in an arrangement 20 for providing an axial scan. Thearrangements 20 and 22 operate as is generally known in the art.

Signals reflected from the means 20 and sample volume 19 back along thereference and sample arms 16, 18 respectively, are coupled back intorespective ports 14 b, 14 c of the splitter 14 and are coupled to adetector 24 which produces axial scan data 26 as is generally known.U.S. Pat. No. 6,341,036, the entire disclosure of which is incorporatedherein by reference, describes systems similar to the one describedabove and shown in FIG. 1.

In general, on scanning the reference arm path length 16, interferencefringes are formed corresponding to positions that match the distance tothe three structures 19 a, 19 b, 19 c in the sample volume 19. Thesingle detector 24 is used to detect the interference fringes. Byenvelope detection of the fringe patterns, an image 26 is constructedthat maps tissue reflectivity to a given location.

As will be apparent from certain exemplary embodiments described hereinbelow, an exemplary embodiment of the present invention relates to asystem which utilizes a detection principle based upon Spectral Radarconcepts (further referred to as Spectral Domain OCT) and/or a hybridmethod between Spectral Domain and Time Domain OCT that is preferablymore sensitive than current state of the art Time Domain OCT, allowing asubstantial increase in the acquisition speed to resolution ratio.

Analysis of the Signal to Noise Ratio (“SNR”) in Time Domain OCT hasbeen previously described in related publications. The interferencefringe peak amplitude in time domain OCT is given by

I _(peak)=√{square root over (P_(ref) p _(sample))},   (1)

with P_(ref), P_(sample) the reference and sample arm power in Watts,respectively. In terms of electrical power at the detector, the signalin units [A²] is defined as

S=η ² e ² P _(ref) P _(sample) /E _(ν) ²,   (2)

with η the quantum efficiency, e the charge quantum and E_(ν)=hc/λ thephoton energy. The reference and sample arm powers are given by therespective reflected spectral densities,

P _(ref,sample) =∫S _(ref sample)(ω)dω  (3)

Assuming that the reference and sample spectral densities are equal tothe source spectral density S(ω), where the sample arm spectral densityis attenuated by a large factor, i.e., S_(ref)(ω)=S(ω),S_(sample)(ω)=αS(ω) with α

1, and inserting the above expression of reference and sample arm intothe original definition of the signal gives,

S=η ² e ² α└∫S(ω)dω┘ ² /E _(ν) ².   (4)

Three contributions to the total noise of OCT signals are: (i) thermalnoise, (ii) shot noise and (iii) relative intensity noise. Thermal noiseis generated by the feedback resistor, shot noise is related to thefinite nature of the charge quantum resulting in statisticalfluctuations on the current, and relative intensity noise is related tothe temporal fluctuations due to chaotic character of classical lightsources. These three contributions to the noise density in units [A²/Hz]are given by,

$\begin{matrix}{{{N_{noise}(f)} = {\frac{4\; {kT}}{R_{fb}} + \frac{2{\eta }^{2}P_{ref}}{E_{v}} + {2\left( \frac{\eta \; {eP}_{ref}}{E_{v}} \right)^{2}\tau_{coh}}}},} & (5)\end{matrix}$

k is Boltzmann's constant, T the temperature in Kelvin, R_(fb) the valueof the feedback resistor, and τ_(coh) the coherence time of the source.Coherence time is related to the full spectral width at half maximum Δλof a Gaussian source by the following relation, τ_(coh)=√{square rootover (2In2/π)}λ₀ ²/(cΔλ). Shot noise limited detection is achieved whenthe second term in Eq. (5) dominates the other noise contributions.

The signal to noise ratio (SNR) is given by

$\begin{matrix}{{{SNR} = \frac{S}{{N_{noise}(f)}{BW}}},} & (6)\end{matrix}$

with BW the signal bandwidth, and parameters S and N_(noise)(f) asdescribed above.Spectral Domain OCT using a Spectrometer and CCD array detector

The best signal to noise performance of Time Domain OCT systems isobtained when the noise is shot noise limited. Shot noise can be reducedsignificantly by replacing the single element detector with amulti-element array detector. When the detection arm light is spectrallydispersed on the array detector, each element of the array detects asmall wavelength fraction of the spectral width of the source. The shotnoise is preferably reduced by a factor equal to the number of elementsof the array. The principle of the signal to noise improvement is basedon the white noise characteristic of shot noise and the observation thatonly electromagnetic waves of the same wavelength produce interferencefringes.

The shot noise power density N_(noise)(f) (in units [W/Hz], [A²/Hz] or[V²/Hz]) is proportional to the current (or equivalently the opticalpower times the quantum efficiency) generated in the detector. For amonochromatic beam of wavelength λ_(I) entering the interferometer, thefringe frequency or carrier f at the detector is determined by thevelocity ν of the mirror, f_(I)=2ν/λ_(I). The shot noise is proportionalto the power (or spectral density S(ω)) at wavelength λ_(I). A secondwavelength λ₂ is preferably coupled into the interferometer. A secondfringe frequency or carrier at frequency f₂=2ν/λ₂ is simultaneouslypresent. The shot noise at this second frequency is preferably the sumof the shot noise generated by the optical power at wavelength λ₁ andλ₂. Also, at frequency f₁ the shot noise is the sum of the shot noisegenerated by the optical power at wavelength λ₁ and λ₂. Thus, at bothfrequencies a cross-shot noise term is generated by the simultaneouspresence of both wavelengths at the detector. By spectrally dispersingeach wavelength to a separate detector, the cross shot noise term can beeliminated. In this way, Spectral Domain OCT offers a significantimprovement of signal to noise ratio over Time Domain OCT systems.

The OCT signal is most easily described in the space domain. For asingle object in the sample arm, the interference term of the OCT signalis proportional to the real part of the Fourier transform of the sourcespectrum S(ω),

I(Δz)oc Re∫exp(ikΔz)S(k)dk,   (7)

with Δz the path length difference between sample and reference arm andk the wave vector. As a function of time, the OCT signal is given by,

I(t)oc Re∫exp(2iωrν/c)S(ω)dω,   (8)

with ν the reference arm mirror velocity. The frequency spectrum of thesignal is given by a Fourier transform of the signal in the time domain,resulting in a complex function. The absolute value of this function isequal to the spectral density,

|I(f)|=|∫I(t)e ^(2 i π∫t) dt|=S(πf c/ν),   (9)

which shows that the signal bandwidth is directly proportional to thesource spectral width and scales linearly with the reference arm mirrorvelocity, i.e., imaging speed. Eq. (9) also preferably directly relatesthe absolute value of the frequency spectrum, |I(f)| to the signal S(see Eq. (4)). Eq. (9) also demonstrates that each angular frequency ofthe light source or equivalently each wavelength of the source isrepresented at its own frequency in the measured interferometric signal.The depth profile information I(t) can be obtained from the complexcross spectral density, |I(f)| by a Fourier transform.

The complex cross spectral density can also be obtained by splitting thesignal I(t) in several spectral bands using a dispersive orinterferometric element. At each detector, only part of the complexcross spectral density is determined. Combining the cross spectraldensities of each detector, the full spectral density of the signal areretrieved. Thus, the same information can be obtained by separatingspectral components to individual detectors. Combining the signal of alldetectors in software or hardware would result in the same signal asobtained with a single detector.

In the detection arm, the spectrum can be split into two equal halves,where two detectors each detect one half of the spectrum. According toEq. (9), the frequency spectra at detectors 1 and 2 are given by|I₁(f)|=S(πfc/ν) for f<f₀, I₁(f)=0 for f>f₀ and I₂(f)=0 for f<f₀,|I₂(f)|=S(πfc/ν) for f>f₀, respectively. The frequency spectrum as wouldbe acquired by a single detector in time domain OCT is given by the sumof I₁(f) and I₂(f); I(f)=I₁(f)+I₂(f). Thus, the signal S after combiningthe spectra is equal, however I₁(f)=0 for f>f₀ and I₂(f)=0 for f<f₀, thebandwidth BW per detector can be reduced by a factor of 2.

The noise is determined by the sum of the shot noise contributions atdetectors one and two. From Eqs. (5) and (6), the shot noise perdetector is proportional to the reference arm power at the detectortimes the bandwidth for the detector. Since the spectrum was split inequal halves, the reference power at detectors 1 and 2 is, respectively,

P_(ref) ¹=0.5P_(ref), P_(ref) ²=0.5P_(ref). (10)

The sum of the shot noise contribution for the two detectors is,

N _(noise) ^(SD) oc P _(ref) ¹×0.5BW+P _(ref) ²×0.5BW=0.5P _(ref) BW,  (11)

which may compared with the shot noise of a single detector in timedomain OCT,

N_(noise) ^(TD) oc P _(ref)BW.   (12)

Thus, by spectrally dispersing the detection and light over two separatedetectors, the signal remains the same, while the noise is reduced by afactor of 2, resulting in a net SNR gain by a factor of 2.

Extending the above analysis, it can be demonstrated that the shot noisecontribution is reduced by a factor equal to the number of detectors.The sum of shot noises for N detector elements, where each detectorelement receives one Nth of the total reference power, is given

$\begin{matrix}{N_{noise} = {\frac{2{\eta }^{2}P_{ref}}{E_{v}}{\frac{BW}{N}.}}} & (13)\end{matrix}$

The signal is the same as in Time Domain OCT, and the SNR ratio forSpectral Domain OCT is given by,

$\begin{matrix}{\frac{S}{N_{noise}} = {\frac{\eta \; P_{sample}N}{2\; E_{v}{BW}}.}} & (14)\end{matrix}$

Thus Spectral Domain OCT enables a SNR improvement over Time Domain OCTof a hundred to a thousand fold, depending on the number of detectorelements N. Using a charge coupled array or an integrating device as adetector, such as, but not limited to, a line scan camera, the ratioN/BCW is replaced by the integration time τ_(i) of the array, whichresults in,

$\begin{matrix}{\frac{S}{N_{noise}} = {\frac{\eta \; P_{sample}\tau_{i}}{2\; E_{v}}.}} & (15)\end{matrix}$

FIG. 2 shows an exemplary Spectral Domain OCT system 100 which includesan interferometer 102 with a source arm 104, a sample arm 106, areference arm 108, and a detection arm 110 with a spectral separatingunit 112, a detector array 114 comprised of a plurality of detectors anda like plurality of amplifiers 116. The amplifiers 116 are coupledthrough optional analog processing electronics (not shown, but known tothose having ordinary skill in the art), and A/D converters (not shown,but known to those skilled in the art) for conversion of signals andthrough digital band pass filtering (“BPF”) units 122 to a processingand display unit 124.

The processing and display unit 124 executes data processing and displaytechniques, and can optionally include the digital band pass filtering(“BPF”) units 122 as well as Digital Fast Fourier Transforms (“DFFTs”)circuits (not shown), in order to provide coherent combination ofsignals and to perform the data processing and display functions. Thedetector array 114 may be 1×N for simple intensity ranging and imagingand/or Doppler sensitive detection, 2×N for dual balanced detection, 2×Nfor simple intensity ranging and/or polarization and/or Dopplersensitive detection, or 4×N for combined dual balanced and polarizationand/or Doppler sensitive detection. Alternatively, an M x N array may beused for arbitrary number “M” of detectors 114 to allow detection oftransverse spatial information on the sample 130.

Electro-magnetic radiation (e.g., light) is transmitted from the sourcealong the source arm 104 to the splitting unit via and is split betweenthe reference arm 108 and the sample arm 106. The light propagates alongthe sample arm to the tissue sample 130 and through the reference arm108 to a wavelength dependent phase arrangement. The light is reflectedfrom the sample and the wavelength dependent phase arrangement backtoward the splitting unit where at least portions of the reflected lightare directed toward the spectral separating unit 112 (which may beprovided as a grating for example). The detection arm light is dispersedby the spectral separating unit 112 and the spectrum is imaged onto thedetector array 114. By stepping the reference arm 108 length over adistance λ/8, the cross spectral density of reference arm 108 and samplearm 106 light can be determined. The processing and display unitreceived the signals fed thereto and performs a Fourier transform of thecross spectral density to generate depth profile information.

FIG. 3A shows a block diagram of an exemplary system according to thepresent invention which illustrates basic principles of a coherentfrequency modulated continuous Wave (“C-FMCW”) system using asingle-frequency tuning source. A monochromatic laser light 70 operableas a frequency chirped laser provides a light signal to an input 72a ofa coupler 72. The coupler 72 divides the light signal into a referencearm 80 which terminates in a reference mirror 82 and a sample arm 84which terminates in a sample 86. The light propagates down paths 80, 84and reflects from the reference mirror 82 and sample mirror 86 toprovide, via coupler 72, interference signals which are detected by aphoto-detector 88.

As shown in graphs of FIGS. 3B-3D, when there is an optical delaybetween two reflected light signals 90 (FIG. 3B) and 92 (FIG. 3C),respectively, a beat signal 94 (see FIG. 3D) having a frequency f may bedetected at the photo detector 88. Where there are multiple reflectionpoints in the sample along the axis, the interference consists of beatnotes having frequencies which are proportional to the optical delaydifference between the reflection (scatter) point in the sample and thereference mirror. The power of each beat frequency component isproportional to the reflectivity of the scatter. Therefore, the image ofthe sample can be constructed by Fourier transform of the interferencedata.

Referring now to FIGS. 4A-4D, in which like elements described above andshown in FIGS. 3A-3D are provided having the same referencedesignations, an optical frequency domain imaging (“OFDI”) systemaccording to an exemplary embodiment of the present invention includes awavelength-swept laser source 95 (also referred to herein as a frequencyswept source 95) which provides a laser output spectrum comprised ofmultiple longitudinal modes to an input of a coupler 72. The coupler 72divides the signal fed thereto into the reference arm 80 whichterminates in the reference mirror 82 and the sample arm 84 whichterminates in the sample 86. The optical signals reflect from thereference mirror 82 and the sample 86 to provide, via coupler 72, aspectrum of signals which are detected by a photo-detector 88.

The center (or mean) wavelength of the signal spectrum is tuned in timeby the creation of new longitudinal modes at the leading side of thespectrum and the annihilation of the modes at the trailing side of thespectrum.

The same principles described above with reference to FIGS. 3A-3D alsoapply to the OFDI technique using a wavelength-swept laser source 95.Similar to the case of a C-FMCW system (e.g., the system of FIG. 3Adescribed above), a beat signal 94 can be produced. In the case of theOFDI system that uses a wavelength-swept laser source, the beat signal94 can be generated having a beat frequency f which corresponds to thedifference in the center frequency of the lights, 96 and 98, from thereference and sample, respectively.

The frequency spacing between longitudinal modes should be substantiallylarger than the detection bandwidth. The mode beat frequency (relativeintensity noise peaks) can be removed by a proper electronic filter,such as low pass filter, prior to digitization. The interference signal94 contains a frequency component that is proportional to the opticaldelay. Furthermore, the image of the sample can be constructed byFourier transform of the digitized interference data.

In one exemplary embodiment of the present invention, thewavelength-swept laser 95 can be provided which utilizes an opticalband-pass scanning filter in the laser cavity to produce a rapidly-sweptmultiple-frequency-mode output. Exemplary filters according to thepresent invention is described below in conjunction with FIGS. 6 and 9A.By using an optical band-pass scanning filter in the laser cavity, it isnot necessary to tune the laser cavity length to provide synchronoustuning of the laser spectrum. Indeed, such arrangement does not requiretuning the longitudinal cavity mode of the laser at the same rate as thecenter wavelength of the laser.

Using the OFDI techniques, a single pixel of the image can beconstructed from the signal that is recorded as a function of time overthe duration of one A-scan through Fourier transform. This is differentfrom the TD OCT where a single pixel is constructed from the datameasured at a short period time within one A-scan. The detectionbandwidth to acquire the same number of data within the same A-scanperiod is approximately the same for both TD and FD OCT. However, theFourier transform used for the OFDI technique effectively improves thesignal-to-noise ratio compared to TD OCT by constructing a single imagepixel from many data points acquired over the whole A-scan period. Thiseffect can result in an “effective” detection bandwidth that is N-timeslarger than the actual detection bandwidth. Therefore, the SNR may beimproved by N times, where N is the number of (digitized) data points inthe Fourier transform. It can be shown that SNR in a shot noise limitedcase is given by:

$\begin{matrix}{\frac{S}{N_{noise}} = \frac{\eta \; P_{sample}N}{2\; E_{v}{BW}}} & (16)\end{matrix}$

Due to the narrowband output spectrum of the wavelength-swept source,however, the relative intensity noise (RIN) can be significantly higherthan that of a CW broadband light source. For a thermal light, RIN isgiven by 1/Δν where Δν=c·λ/λ² is the optical bandwidth of the(instantaneous) source output. For a laser light, RIN results fromdifferent statistics and therefore has a different value than thethermal light. For FD-OCT, a wavelength-swept laser with a low RIN levelis preferred. The laser light with multiple longitudinal modes may havea similar RIN level as the thermal light with the same linewidth. Inthis case, a means to suppress the RIN is critical to have sufficientSNR, such as the dual balanced detection.

Use of a swept source results in a system having reduced shot noise andother forms of noise which allows for much lower source powers, or muchhigher acquisition rates than current systems. The increased detectionsensitivity allows for real time imaging. Such imaging speed can assistwith a problem of motion artifacts, such as in gastrointestinal,ophthalmic and arterial imaging environments. By increasing the framerate while maintaining or improving the signal to noise ratio suchartifacts can be minimized. The present invention also enables one toscreen large areas of tissues with the OFDI technique, and allowsclinical viable screening protocols using this method.

For ophthalmic applications of OFDI, the efficient detection preferablyallows for a significant increase of acquisition speed. A possiblelimitation in ophthalmic applications is the power that is allowed toenter the eye according to the ANSI standards (approximately 700microwatts at 830 nm). Current data acquisition speed in ophthalmicapplications is approximately 100-500 A-lines per second. The powerefficient detection would allow for A-line acquisition rates on theorder of about 100,000 A-lines per second, or video rate imaging atabout 3,000 A-lines per image.

The gain in SNR is achieved because the shot noise has a white noisespectrum. The signal intensity present at the detector at frequency w(or wavelength λ) contributes only to the signal at frequency ω, but theshot noise is generated at all frequencies. By narrowing the opticalband width per detector, the shot noise contribution at each frequencycan be reduced, while the signal component remains the same.

FIG. 5 shows an exemplary embodiment of a system 99 for performingoptical imaging using frequency-domain interferometry (“OFDI”) whichincludes a frequency swept source 100 that emits a narrowband spectrumof which the center wavelength is tuned continuously and repeatedly intime across the bandwidth of the gain medium in the source. Theinstantaneous emission spectrum consists of a plurality of frequencymodes of the light source. The frequency swept source 100, may beprovided in a variety of different ways, some of which are describedbelow. The source 100 may, for example, be provided from various gainmedia, tunable wavelength filters, cavity configurations. Devices andmethods are known in the art to provide a rapidly-tuned wavelength-sweptlaser source, such as solid-state lasers, active-ion-doped waveguidelasers, and fiber lasers. A wavelength-swept laser in a mode-lockedregime can also be used with potential advantage of a lower relativeintensity noise (RIN) in a frequency region between harmonics oflongitudinal-mode beat frequencies. An optical saturable absorber may beincorporated inside a laser cavity or after the output port of thesource to lower RIN level.

The light provided from swept source 100 is directed toward afiber-optic coupler 102 which divides the light fed thereto into areference arm 103 and a sample arm 104. In this exemplary embodiment,the coupler 102 has a 90:10 power splitting ratio with 90% of the powerbeing directed toward the sample arm. Those of ordinary skill in the artwould understand, however, that other coupling ratios for the coupler102 may also be used. The particular coupling ratio to use in anyparticular application should be selected such that an amount of poweris provided to both the reference arm and the sample arm to allow forproper operation of the exemplary system according to the presentinvention.

The power provided to the sample arm passes though a circulator 111, andilluminates a sample 136 to be imaged through a transverse-scanningimaging probe. The reference arm provides preferably a fixed opticaldelay. The lights reflected from a reference mirror 124 and from withinthe sample 136 can be directed through the respective circulators 110,111 toward a fiber-optic beam splitter (or fused coupler) 150 andinterfere between each other to produce interference signals.

It is desirable that the combining coupler 150 have an equal splittingratio with minimal polarization dependence and wavelength dependenceover the wavelength tuning range of the source. A deviation from equalsplitting results in reduction of common mode rejection ratio (“CMRR”)of the dual balanced detection. In one embodiment, the combining coupler150 is preferably provided as a bulk broadband beam splitter. Those ofordinary skill in the art would understand that other types of couplers(including but not limited to wavelength-flattened fiber fused couplers)may also be used.

The interference signals are received by a dual balanced receiver 151.Output of the receiver 151 is provided to a computing arrangement (e.g.,a data acquisition board and computer 160), such that the output isdigitized and processed by the computer arrangement to produce an image.The data acquisition, transverse scanning, and wavelength tuning aresynchronously controlled.

FIG. 6 shows an exemplary light source 100′ which may, for example, beadapted for use as a frequency swept source (such as frequency sweptsource 100 described above with reference to FIG. 5) is provided from anoptical filter 170, coupled through a lens 172 and a light path 174 to alight source/controller 176 (hereinafter referred to as “lightcontroller 176”). The light controller 176 may, in turn, be coupled toone or more applications 178. The applications 178 may, for example,correspond to optical imaging processes and/or optical imaging systems,laser machining processes and systems, photolithography andphotolithographic systems, laser topography systems, telecommunicationsprocesses and systems. Thus, the exemplary light source 100′ providedfrom the filter 170 and the light controller 176 may be used in a widevariety of different applications, certain general examples of which aredescribed herein.

As shall be described in further detail below, the filter 170 allows thelight source 100′ to operate as a frequency swept source which emits aspectrum of which the center wavelength can be tuned continuously andrepeatedly in time across the bandwidth of the light controller 176.Thus, light source 100′ may have an instantaneous emission spectrumcomprised of a plurality of frequency modes of the lightsource/controller 176. In this exemplary embodiment, the opticalwavelength filter 170 is configured as a reflection-type filter in thatthe input and output ports are identical. Thus, light path 174 may beprovided, for example, as an input/output optical fiber and lens 172 maycorrespond to a collimating lens. Although the filter 170 in FIG. 6 isshown coupled to one or all of applications 178 through the lightcontroller 176, it is possible to directly couple the filter 170 to oneor more of the applications 178. Alternatively, it is possible to couplethe filter 170 to one or more of the applications 178 through a deviceother than a light controller.

In the exemplary embodiment according to the present invention, thelight controller 176 can include a number of systems that arespecifically adapted to transmit a beam of light (in one embodiment, acollimated beam of light) having a broad frequency (f) spectrum. Inparticular, the beam of light can include a plurality of wavelengths,within the visible light spectrum (e.g., red, blue, green). The beam oflight provided by the light controller can also include a plurality ofwavelengths that are defined outside of the visible spectrum (e.g.,infrared).

As shall be described in greater detail below with reference to FIG. 7,in one exemplary embodiment of the present invention, the lightcontroller 176 can include a unidirectional light transmission ring. Inanother exemplary embodiment to be described in detail in conjunctionwith FIG. 9 below, the light controller 176 can include a linearresonator system. The filter 170 includes a wavelength dispersingelement 180 adapted to receive the beam of light from the lightcontroller 176 and to separate the beam of light into a plurality ofdifferent wavelengths of light each directed along a light path as isknown. The wavelength dispersing element 180 can include one or moreelements adapted to receive the beam of light from the light controller176, and to separate the beam of light into a plurality of wavelengthsof light each directed along a light path. The wavelength dispersingelement 180 is further operative to direct the plurality of wavelengthsof light in a plurality of angular directions or displacements withrespect to an optical axis 182. In one exemplary embodiment of thepresent invention, the wavelength dispersing element 180 can include alight dispersion element, such as a reflection grating 184. Thewavelength dispersing element 180 could alternatively be provided as atransmission grating (e.g. a transmission type grating such asDickson-type holographic grating), a prism, a diffraction grating, anacousto-optic diffraction cell or combinations of one or more of theseelements.

The wavelength dispersing element 180 directs light at each wavelengthtowards a lens system 186 along paths which are at an angle with respectto the optical axis 182. Each angle is determined by the wavelengthdispersing element 180. The lens system 186 can include one or moreoptical elements adapted to receive the separated wavelengths of lightfrom the wavelength dispersing element 180 and to direct or steer and/orfocus the wavelengths of light to a predetermined position located on abeam deflection device 188. The beam deflection device 188 can becontrolled to receive and selectively redirect one or more discretewavelengths of light back along the optical axis 182 through the lenssystem 186 to the wavelength dispersing element 180 and back to thelight controller 176. Thereafter, the light controller 176 canselectively direct the received discrete wavelengths of light to anyoneor more of the applications 178. The beam deflecting device 188 can beformed and/or arranged in a number of ways. For example, the beamdeflecting device 188 can be provided from elements including, but notlimited to, a polygonal mirror, a planar minor disposed on a rotatingshaft, a minor disposed on a galvanometer, or an acousto-opticmodulator.

In the exemplary embodiment shown in FIG. 6, the dispersing element 186includes a diffraction grating 184, a lens system 186 (which has firstand second lenses 190, 192 to form a telescope 193), and the beamdeflecting device 188 which is shown as a polygon mirror scanner 194.The telescope 193 is provided from the first and second lenses 190, 192with 4-f configuration. The first and second lenses 190, 192 of thetelescope 193 are each substantially centered on the optical axis 182.The first lens 190 is located at a first distance from the wavelengthdispensing element 180 (e.g., diffraction grating 184), which isapproximately equal to the focal length F1 of the first lens 190. Thesecond lens 192 is located a second distance from the first lens 190,which is approximately equal to the sum of the focal length F1 of thefirst lens 190 and the focal length F2 of the second lens 192. In thisexemplary arrangement, the first lens 190 can receive the collimateddiscrete wavelengths of light from the wavelength dispersing element180, and may effectively perform a Fourier Transform on each one of thecollimated one or more discrete wavelengths of light to provide an equalone or more converging beams projected onto an image plane (seedesignated IP of FIG. 6). The image plane IP is located between thefirst and second lenses and at a predetermined distance from the firstlens, which predetermined distance is defined by the focal length F1 ofthe first lens. After propagating through the image plane IP, theconverging beam(s) form an equal one or more diverging beams that arereceived by the second lens. The second lens operates to receive the oneor more diverging beams and to provide an equal number of collimatedbeams having predetermined angular displacements with respect to theoptical axis 182 for directing or steering the collimated beams topredefined portions of the beam deflection device 188.

The telescope 193 is configured to provide a number of features, asdescribed above, and further to convert diverging angular dispersionfrom the grating into converging angular dispersion after the secondlens 192, which is desired for proper operation of the filter 170. Inaddition, the telescope 193 provides a useful degree of freedom, whichcontrols the tuning range and reduces the beam size at the polygonmirror 194 to avoid a beam clipping.

As is illustrated in FIG. 6, the polygon mirror 194 reflects backpreferably only the spectral component within a narrow passband as afunction of the angle of the front mirror facet of the polygon withrespect to the optic axis. The reflected narrowband light is diffractedand received by the optical fiber 174.

The orientation of the incident beam with respect to the optic axis andthe rotation direction 198 of the polygon mirror 194 determine thedirection of wavelength tuning wavelength up (positive) scan or down(negative) scan. The arrangement in FIG. 6 produces a positivewavelength sweep. It should be understood that while the mirror 194 isshown in FIG. 6 as having twelve facets, fewer or more than twelvefacets can also be used. The particular number of facets to use in anyapplication depends upon the desired scanning rate and scanning rangefor a particular application. Furthermore, the size of the mirror isselected in accordance with the needs of a particular application takinginto account factors including, but not limited to, manufacturabilityand weight of the mirror 194. It should also be appreciated that lenses190, 192 may be provided having different focal lengths. The lenses 190,192 should be selected to provide a focal point at approximately thecenter point 200 of the mirror 194.

Consider a Gaussian beam with a broad optical spectrum incident to thegrating from the fiber collimator 172. The well-known grating equationis expressed as λ=p·(sin α+sin β) where λ is the optical wavelength, pis the grating pitch, and α and β are the incident and diffracted anglesof the beam with respect to a nominal axis 202 of the grating,respectively. The center wavelength of tuning range of the filter isgiven by λ₀=p·(sin α+sin β₀) where λ₀ is the angle between the opticaxis 38 of the telescope and the grating normal axis. It can be shownthat FWHM bandwidth of the filter is given by (δλ)_(FWHM)/λ₀=A·(p/m)cosα/W where A=√{square root over (4 In2/π)} for double pass, m is thediffraction order, and W is 1/e²-width of the Gaussian been at the fibercollimator. When the real part of the complex spectral density isdetermined, ranging depth z is defined by

$z = {\frac{\lambda_{0}^{2}}{4({\delta\lambda})_{FWHM}}.}$

Tuning range of the filter is fundamentally limited by the finitenumerical aperture of lens 1 20. The acceptance angle of lens1 withoutbeam clipping is given by Δβ=(D₁ −W cos β ₀/cos α)/F₁, where D₁ and F₁are the diameter and focal length of lens1. It relates to the filtertuning range via Δλ=p cos β₀·Δβ. An importance design parameter of thefilter, originated from multiple facet nature of the polygon mirror, isthe free spectral range, which is described in the following. A spectralcomponent after propagating through lens1 20 and lens2 22 will have abeam propagation axis at an angle β′ with respect to the optic axis 38:β′=−(β−β₀)·(F₁/F₂) where F₁ and F₂ are the focal lengths of lens1 andlens2, respectively. The polygon has a facet-to-facet polar angle givenby θ=2π/N≈L/R where L is the facet width, R is the radius of thepolygon, and N is the number of facets. If the range of β′ of incidentspectrum is greater than the facet angle, i.e. Δβ′=Δβ·(F₁/F₂)>θ, thepolygon minor could retro-reflect more than one spectral component at agiven time. The spacing of the multiple spectral componentssimultaneously reflected, or the free spectral range, can be shown to be(Δλ)_(FSR)=p cos β₀(F₂/F₁)·θ.

In the application as an intracavity scanning filter, the tuning rangeof the laser cannot exceed the free spectral range if the gain mediumhas homogenous broadening, since the laser chooses the wavelength ofhighest gain. The duty cycle of laser tuning by the filter can be, inprinciple, 100% with no excess loss caused by beam clipping if twonecessary conditions are met as follows:

$\begin{matrix}{W < {\frac{\cos \; \alpha \; F_{1}}{\cos \; \beta_{0}\; F_{2}}L\mspace{14mu} {and}\mspace{14mu} W} < {\frac{\cos \; \alpha}{\cos \; \beta_{0}}{\left( {F_{2} - S} \right) \cdot \theta}}} & (17)\end{matrix}$

The first equation is derived from the condition that the beamwidthafter lens 192 should be smaller than the facet width. The secondequation is from that the two beams at the lowest and highestwavelengths 204, 206 respectively of the tuning range should not overlapeach other at the polygon mirror S in Equation (1) denotes the distancebetween the lens 192 and the front mirror of the polygon.

In one experiment, optical components with the following parameters wereselected: W=1.9 mm, p=1/1200 mm, α=1.2 rad, β₀=0.71 rad, m=1, D₁=D₂=25mm, F₁=100 mm, F₂=45 mm, N=24, R=25 mm, L=6.54, S=5 mm, θ=0.26 rad,λ₀=1320 nm. From the parameters, the theoretical FWHM bandwidth, tuningrange and free spectral range of the filter could be calculated:(δλ)_(FWHM)=0.09 nm, Δλ=126 nm and (Δλ)_(FSR)=74 nm. Both conditions in(1) are satisfied with margins. The characteristics of the filter weremeasured using broadband amplifier spontaneous emission light from asemiconductor optical amplifier (SOA) and an optical spectrum analyzer.The optical spectrum analyzer recorded the nonnalized throughput(reflected) spectrum in peak-hold mode while the polygon mirror wasspinning at its maximum speed of 15.7 kHz. The measured tuning range was90 nm which is substantially smaller than the theoretical value of 126nm. The discrepancy was due to the aberration of the telescope,primarily field curvature, associated with relatively large angulardivergence of the beam from the grating. It is expected that theaberration would be improved by using optimized lenses. The freespectral range was 73.5 nm in agreement with the theoreticalcalculation. The FWHM bandwidth was measured to be 0.12 nm. Thediscrepancy with theoretical limit of 0.11 nm may be reasonableconsidering the aberration and imperfection of the optical elements.

FIG. 7 shows an extended-cavity semiconductor laser 208 according to anexemplary embodiment of the present invention which can include a filter210 that may, for example, be similar to the filter 170 described abovewith reference to FIG. 6. The filter 210 is coupled through a lightdirecting element 212 and a light path 214 to a Faraday circulator 216.In this exemplary embodiment, the filter 210 includes a grating 232 anda polygonal mirror 236. Thus, the filter 210 may correspond to apolygon-based filter. A motor 234 drives the mirror.

The Faraday circulator 216 of this exemplary embodiment is coupledthrough polarization controllers 220, 222 to a gain medium 224 which inone exemplary embodiment can be a semiconductor optical amplifier (e.g.,SOA, Philips, CQF 882/e) having coupled thereto a current source 226which provides an injection current to the SOA 224. The intracavityelements may be connected by single-mode optical fibers, for example.The two polarization controllers 220, 222 can align the polarizationstates of the intracavity light to align to the axes of maximumefficiency of the grating 232 and of maximum gain of the SOA 224.

A laser output 228 may be obtained through a 90% port of a fiber-opticfused coupler 230. To generate a sync signal useful for potentialapplications, 5% of the laser output may be coupled through a variablewavelength filter 237 having a bandwidth of 0.12 nm and is directedtoward a photodetector 238. In one exemplary embodiment, the centerwavelength of the filter may be fixed at 1290 nm. The detector signalgenerates short pulses when the output wavelength of the laser is sweptthrough the narrowband passband of the fixed-wavelength filter. Thetiming of the sync pulse is controlled by changing the center wavelengthof the filter.

FIG. 8A shows a graph 240 of an output spectrum of a laser of the typedescribed above with reference to FIG. 7 as measured by an opticalspectrum analyzer in peak-hold mode, when the polygon mirror (i.e.mirror 236 in FIG. 7) spins at a rate of 15.7 kHz. The edge-to-edgesweep range may be from 1282 nm to 1355 nm over 73 nm-width equal to thefree-spectral range of the filter. The Gaussian-like profile of themeasured spectrum, rather than a square profile, is likely due to thepolarization-dependent cavity loss caused by polarization sensitivity ofthe filter and the birefringence in the cavity. It is preferable toadjust the polarization controllers to obtain the maximum sweep rangeand output power.

FIG. 8B shows a curve 242 of a laser output in a time domain. The uppertrace 244 corresponds to a sync signal obtained through thefixed-wavelength filter. The amplitude of power variation from facet tofacet was less than 3.5%. The peak and average output power was 9 mW and6 mW, respectively. It should be mentioned that the y-axis scale ofcurve 240 had to be calibrated from the time-domain measurement, becausethe optical spectrum analyzer only recorded a time-averaged spectrum dueto the laser tuning speed much faster than the sweep speed of thespectrum analyzer. A frequency downshift in the optical spectrum of theintracavity laser light may arise as the light passes through an SOAgain medium (e.g. SOA 224 in FIG. 7), as a result of intraband four-wavemixing phenomenon. In the presence of the frequency downshift, thepositive wavelength scan can facilitate tuning of the laser spectrum,and thereby produce higher output powers. The peak power of the laseroutput can be measured as a function of the tuning speed. The negativetuning speed may be obtained by flipping the position of the collimatorand the orientation of the grating with respect to an optic axis (e.g.,axis 182 in FIG. 6). It is preferable to make the physical parameters ofthe filter approximately identical in both tuning directions. Thus, thecombined action of self-frequency shift and positive tuning allowshigher output to be obtained and enables the laser to be operated athigher tuning speed. Therefore, the positive wavelength scan may be thepreferred operation. The output power may decrease with increasingtuning speed. Thus, a short cavity length may be desired to reduce thesensitivity of the output power to the tuning speed. In this case, afree-space laser cavity is preferred.

FIG. 9A shows an exemplary embodiment of a free-space extended-cavitysemiconductor tunable laser 250 according to an exemplary embodiment ofthe present invention that includes a semiconductor waveguide 252fabricated on a substrate chip 254 coupled to a polygon scanning filter255 through a collimating lens 256. A front facet 258 may beanti-reflection coated, and an output facet 260 is cleaved or preferablycoated with dielectrics to have an optimal reflectivity. An output 262of the laser is obtained through the output coupling lens 264. Thecollimating lenses 256, 264 are preferably provided as aspheric lenses.

The filter 255 includes a wavelength dispersing element 180′ adapted toreceive the beam directed thereto from the lens 256. The wavelengthdispersing element 180′ may be similar to wavelength dispersing element180 described above with reference to FIG. 6. A lens system 186′ can bedisposed between the wavelength dispersing element 180′ and a beamdeflection device 188′. The wavelength dispersing element 180′ and abeam deflection device 188′ may be similar to wavelength dispersingelement 180 and a beam deflection device 188 described above withreference to FIG. 6. The lens systems 186′ includes a pair of lenses 187a, 187 b which are preferably provided as achromats having lowaberration particularly in field curvature and coma.

A sync output may be obtained by using a lens 266, a pinhole 268, and aphotodetector 270 positioned on the 0-th order diffraction path for thelight which is on retro-reflection from a polygon scanner 272. Thedetector generates a short pulse when the focus of the optical beam of aparticular wavelength sweeps through the pinhole 268. Other types ofgain medium may include, but are not limited to, rare-earth-ion dopedfiber, Ti:Al₂O₃, and Cr⁴⁺:forsterite.

FIG. 9B, shows another exemplary embodiment of a wavelength tunablefilter 280 according to the present invention which may include anoptical fiber 281 coupled to an input collimating lens 282, opticallycoupled to a diffraction grating 284, a focusing lens 286, and aspinning disk 288. The diffraction grating 284 may be replaced by otherangular dispersive elements such as a prism. In one exemplaryembodiment, the diffraction grating 284 can have a concave curvaturewith a focal length selected such the focusing lens 286 is not needed.

Preferably more than one reflector 290 may be deposited on a surface288a of the spinning disk 288. Preferably, the reflectors 290 comprisemultiple narrow stripes periodically and radially patterned. Thematerial for the reflectors is preferably gold. The disk 288 can becomposed of a lightweight plastic or silicon substrate. Instead of thereflectors deposited on the top surface of the disk, the disk can have aseries of through holes followed by a single reflector attached to theback surface of the disk. Incident from the optical fiber 281, theoptical beams of different wavelengths may be illuminated on the surfaceof the disk into a line after being diffracted by the grating andfocused by the lens 286 (in those systems which include lens 286).Preferably, only the beam that impacts the reflectors of the spinningdisk may be retro-reflected and received by the optical fiber 281. Amirror 292 may be used to facilitate the access of the beam onto thedisk.

The distance from the lens 286 to the reflectors of the disk 288 isequal to the focal length, F, of the lens. It can be shown from thegrating equation that the tuning range of the filter is given by Δλ=pcos β₀(D/F) where D denotes the distance between the stripes. The widthof the strip, w, is preferably made to be substantially equal to thebeam spot size w_(s), at the surface of the disk:

$w_{S} = {W{\frac{\cos \; \beta_{0}}{\cos \; \alpha} \cdot \frac{F/z}{\sqrt{1 + \left( {f/z} \right)^{2}}}}}$

where z=πw_(s) ²/λ. It leads to a FWHM filter bandwidth given by(δλ)_(FWHM)/λ₀=A·(p/m)cos α/W where A=√{square root over (4 In 2)}/π.For the w>w_(s), filter bandwidth becomes greater, and for w<w_(s), theefficiency (reflectivity) of the filter is decreased by beam clipping.The orientation of an incident beam 294 with respect to the optic axisof the lens 286 and the spinning direction 288 preferably determines thesense of wavelength tuning The positive wavelength scan is preferable,which is achieved by spinning the disk 288 in a clockwise direction asshown in FIG. 9B.a. Interferometer

FIG. 10A shows an exemplary embodiment of an OFDI system 300 accordingto the present invention for performing optical imaging usingfrequency-domain interferometry includes a frequency swept source 301which emits a light signal having an instantaneous emission spectrumcomprised of a plurality of frequency modes of the light source. Source301 may, for example, be provided as one of the sources described abovewith reference to FIGS. 4A, 5, 6, 7, 9 and 9B. The light from source 301can be directed toward a fiber-optic coupler 302 which divides the lightfed thereto into a reference arm 303 and a sample arm 304.

The reference arm 303 preferably includes a polarization circuit 306 anda circulator 308. Thus, light propagates from source 301 through thecoupler 302, the polarization circuit 306 and the circulator 308 to anoptional motion artifact circuit 309. The optional motion artifactcircuit 309 may be provided from a lens 310 which directs the lighttoward a frequency shifter 311, a phase tracker 312 and a dispersioncompensator 314. The light passes through optional circuit 309 and isincident upon a reference mirror 316. It should be appreciated thatcircuit 309 functions to remove or reduce motion artifacts. It shouldalso be appreciated that circuit 309 may include all of the elements310-314, and/or one or more of the circuit elements 310-314.

The sample arm 304 may include a circulator 318. Thus, a light signaltransmitted from the source 301 propagates from source 301 through thecoupler 302 and the circulator 308 to a lens 320 which directs the lighttoward a scanning mirror 322. The scanning mirror 322 may be providedfrom a wide variety of optical elements including but not limited to, agalvanometer, a piezoelectric actuator or another functionallyequivalent device. A transverse scanner 324 is coupled to the scanningmirror 322 and a data acquisition board and computer 326. The dataacquisition board and computer 326 is also coupled to the frequencyswept source 301.

The OFDI system 300 shown in FIG. 10A can also include a polarizationdiversity balanced detection (“PDBD”) circuit 334 configured to receivesignals from the reference arm 303 and/or the sample arm 304. Inparticular, the reference arm 303 is connected through circulator 308and polarization control circuit 330 to a reference port of the PDBDcircuit 334. Similarly, sample arm 304 is connected through circulator318 and polarization control circuit 332 to a sample port of the PDBDcircuit 334.

b. Interferometer

The sample arm 304 collects light reflected from a tissue sample 328 andis combined with the light from the reference arm 303 in thepolarization diversity balanced detection (PDBD) circuit 334 to forminterference fringes.

For example, the OFDI technique does not require that the optical pathlength in the reference arm be scanned in time. Thus, in certainexemplary embodiments of the present invention, it may be preferable toprovide the reference arm as a fixed delay reference arm. Such fixeddelay reference arms may have various configurations that are known tothose having ordinary skill in the art.

The reference arm 303 can be either of reflective and/or transmissiontype, and can return light back from the mirror 316. The returned lightis directed toward the polarization control circuit 330 via thecirculator 308. Similarly, the reflected light from the sample 338 canbe directed toward a polarization control circuit 332 via the circulator318. The reference arm can also be transmission with no reflection. Thepolarization control circuit 330 can be used to match the polarizationstate of the reference-arm light to that of the sample-arm. The totalbirefringence in the interferometer should be minimized not to inducewavelength-dependent birefringence. The polarization controller mayinclude, but is not limited to, a fiber-optic polarization controllerbased on bending-induced birefringence or squeezing.

Preferably, the chromatic dispersion should be matched substantiallybetween the reference and sample arm. The result of strong dispersionmismatch may be a loss in the axial resolution. Any residual dispersioncan likely be compensated by appropriate signal processing, such asnonlinear mapping based on interpolation of the detector data before theFourier transform. This mapping may also be accomplished, at least inpart, by adjusting the optical layout of the wavelength-swept source. Inone example in which the source 301 includes a polygon scanner and atelescope, the distance between the polygon scanner and the telescopecan be adjusted to convert wavelength space to wave vector space priorto Fourier transformation.

c. Sample Arm

For certain OFDI applications, the sample arm may be terminated by anoptical probe comprising a cleaved (angled, flat, or polished) opticalfiber or free space beam. A lens 336 (such as, but not limited to,aspherical, gradient index, spherical, diffractive, ball, drum or thelike) may be used to focus the beam on or within the sample. Beamdirecting elements (such as, but not limited to, mirror, prism,diffractive optical element or the like) may also be contained withinthe probe to direct the focused beam to a desired position on thesample. The position of the beam may be changed on the sample as afunction of time, allowing reconstruction of a two-dimensional image.Altering the position of the focused beam on the sample may beaccomplished by the scanning mirror 322. The scanning mirror 322 may beprovided, for example, from a number of different devices including, butnot limited to, a galvanometer, piezoelectric actuator, an electro-opticactuator or the like.

The sample arm probe may be a fiber optic probe that has an internallymoving element, such that the motion is initiated at a proximal end ofthe probe and the motion is conveyed by a motion transducing arrangement(such as, but not limited to, wire, guidewire, speedometer cable,spring, optical fiber and the like) to the distal end. The fiber opticprobe may be enclosed in a stationary sheath which is opticallytransparent where the light exits the probe at the distal end. Thus,scanning way also be accomplished by moving the optical fiber. Forexample, by rotating the optical fiber, or linearly translating theoptical fiber. FIG. 10B shows an exemplary embodiment of the probe 359which includes an inner cable 361 (that may rotate or linearly translatealong the axis of the probe), an outer transparent or semi-transparentsheath 362, distal optics 364, and remitted light 366 (which may be atany angle with respect to axis of catheter).

d. Detection

The PDBD circuit 334 may include a plurality of detectors 370 disposedto provide dual balanced detection. Dual balanced detection may bepreferred in certain applications for the following reasons. First, mostlight sources generate 1/f noise (f=frequency) at relatively lowfrequencies and balanced detection will eliminate 1/f source noise.Second, an interference term of the sample arm light with itself (i.e.an auto-correlation term) can be present on top of the true signal term,which is preferably the interference between sample and reference arm.Such auto-correlation term can be eliminated by a differential techniqueand balanced detection may eliminate this auto-correlation term from themeasured signal. Third, RIN can be reduced.

The detectors 370 may preferably include photodiodes (such as, but notlimited to, silicon, InGaAs, extended InGaAs, and the like). Balanceddetection can be implemented by subtracting diode signals that areexactly out of phase with respect to the maxima and minima pattern. Thedifference between two detector signals is obtained by a differentialcircuit included in PDBD circuit 334 and amplified by trans-impedanceamplifiers (“TIA”) 360. The dual balanced receiver may be furtherfollowed by a low-pass or band-pass filter to reject noise outside thedetection bandwidth.

In this exemplary embodiment of the present invention, the dual balanceddetection can be implemented as follows. The polarization beam splitter362 receives signals from the reference and sample arms and provides twooutput signals. The two output signals are further split by twonon-polarizing beam splitters 364 a, 364 b, respectively. The outputsfrom each beam splitter 364 a, 364 b are detected by a dual balancedreceiver provided from the four detectors 370. Furthermore, the twooutputs of the dual balanced receivers are digitized and processed in acomputer arrangement to obtain a polarization diversity.

The receiver output is provided to circuit 326 which acquires anddigitizes the signals fed thereto via A/D converters, and stores thedigitized signals in a computer for further processing. The bandwidth ofthe TIA is preferably matched to half the sampling rate. Gain of the TIAis preferably selected such that the maximum receiver output range ismatched to the voltage range of the A/D converter.

e. Processing

If more than two detectors are used, the signals can be selectivelysubtracted and complex spectral density can be obtained. Using theFourier transform, the complex cross spectral density can be convertedto a depth profile in the tissue. Several methods to process the complexspectral density to obtain depth profile information are known to thoseskilled in the art, such as, but not limited to, by obtaining at leasttwo signals with a Pi/2 phase shift in the reference arm and thenreconnecting the complex spectral density by some linear combination ofthe two signals, or by squaring the spectral density.

Following the detection, analog processing can include a trans-impedanceamplifier, low pass (band pass) filter, and digitization of the signal.This signal may then be converted to reflectivity as a function of depthby the Fourier transform operation. Digital processing includesdigitization, digital band pass filtering in either the frequency domainor time domain (FIR or IIR filter) and inverse Fourier transformation torecover the tissue reflectivity as a function of depth.

Prior to the Fourier transformation, the detected non-linear wavelengthcoordinates is preferably converted to regularly sampled wave-vectorspace. Typically zero padding the signal, Fourier transformation, andinverse Fourier transformation with re-sampling can be utilized forremapping. Other interpolation methods known in the art, such as linear,bi-linear, and cubic spline interpolation of the data may also be usedto convert wavelength space into regularly sampled k space. This mappingmay also be accomplished in part by adjusting the optical layout of thewavelength-swept source. In one example, the distance between thepolygon scanner and the telescope may be adjusted to convert wavelengthspace to wavevector space prior to Fourier transformation.

Another exemplary embodiment of the present invention can utilize one ormore techniques described below to further enhance the performance andfunctionality of imaging. These techniques are not limited to the OFDItechniques that use a multiple-frequency-mode tuned source, but can beapplied in the OFDI technique using a single-frequency tuned source.

a. Polarization Diversity

For an application where polarization fading is a problem, apolarization diversity scheme may be used. Various configurations forpolarization diversity are known in the art.

In the system shown in FIG. 10A, the polarization diversity circuitoperates as follows. The polarization beam splitter 362 separates thereference-arm and sample-arm light signals depending upon theirpolarization states. The polarization controller 330 is preferablyadjusted so that the reference-arm power is split with an equalmagnitude by the polarization controller. The polarization state of thesample arm power can be assumed to vary randomly due to the probe orsample motion, therefore the separating ratio of the sample arm power bythe polarization splitter can vary in time. However, the two outputsignals at the two output ports of the polarization beam splitter 362can be detected by a photo receiver, e.g., squared and summed. Theresulting signal is independent of the polarization state of the samplearm light.

b. Carrier-Frequency Heterodyne Detection

The optical frequency shifter 311 may be situated in the reference arm303 to shift the optical frequency for carrier-frequency heterodynedetection. As a result, the signal frequency band is shifted by themagnitude of the frequency shift. In this manner, relatively large 1/fnoise (f=frequency) and RIN around DC can be avoided. The frequencyshifter can be, but not limited to, an acousto-optic frequency shifter.In the detection, a proper electronics should be used to demodulate thecarrier frequency.

One of the benefits of using the frequency shifter 311 is that theeffective ranging depth can be doubled. This can be illustrated in theelectrical frequency domain, as shown in FIG. 10C in which a graph 380depicts the fringe visibility curve given by the instantaneous outputspectrum of the source. The visibility curve has a Gaussian profile ifthe source's instantaneous spectrum is Gaussian. A curve 390 depicts thetransmission efficiency profile of an electrical filter, which isoptimized for a given Nyquist frequency defined as the half of thesampling frequency. Section (a) of FIG. 10C shows a typical case wherethere is no frequency shifter in the interferometer and the electricalfilter is a low pass filter. Because the positive and negative frequencyband is not differentiable, the images associated with the positive andnegative frequency band, respectively, are overlapped. Because of thisambiguity, only half of the frequency range (zero to f_(N)) or (zero to−f_(N)) is usable in this case. However, using a frequency shifterresults in a shift of the visibility curve by f_(FS), as shown inportion (b) of FIG. 10C. With a bandpass filter (or a low pass filter),both sides of the frequency band centered at f_(FS) produce imageswithout ambiguity, resulting in a twice larger ranging depth compared tosection (a) of FIG. 10C.

Instead of a square-top bandpass filter, it is possible to use a slopefilter. In an example shown in FIG. 10C section (c), the transmissionefficiency curve of the filter, 390, has an exponentially-rising(falling) slope in its low frequency band. This filter may be useful inwhich attenuation is significant and the resulting signal strengthdecays with depth. The slope filter can improve the dynamic range of thedetection by effectively suppressing the large signal from the surfacerelative to that at greater depths.

. Reference Arm Delay (Phase Tracking and Auto-Ranging)

As described above, the OFDI technique does not: require the opticalpath length in the reference arm to be scanned in time. A fixed-delayreference arm can be made in various configurations that are known tothose having ordinary skill in the art. The reference arm can be ofeither reflective or transmission type.

In certain applications, the capability of varying the optical delay inthe reference arm may be useful when a larger ranging depth is desired,without increasing the data acquisition rate or reducing theinstantaneous linewidth of the optical source. Such ability is useful ina clinical study where the distance from the imaging lens and the frontsurface of the sample can varies significantly. Such variation canresult from the motion or from the uncontrolled position of a probingcatheter. For example, a rotating catheter inside a blood vessel canhave distance variation by a couple of millimeter over a single A-scan.

A mechanism in the reference arm 303 may allow for scanning the groupdelay of the reference arm 303. This group delay can be produced by anyof a number of techniques known to those having ordinary skill in theart, such as, but not limited to, stretching an optical fiber, freespace translational scanning using a piezoelectric transducer, or via agrating based pulse shaping optical delay line. Preferably, the delaycan be introduced by a non-mechanical or motionless arrangement. By theterm “non-mechanical”, what is meant is that there are no mechanicallymoving parts being utilized. The absence of the mechanically movingparts is believed to reduce the known deficiencies of using mechanicaldevices to introduce delay. In contrast to traditional LCI or OCTsystems, the reference arm 303 according to an exemplary embodiment ofthe present invention does not necessarily need to scan over the fullranging depth in the sample, and can preferably scan over at least afraction of the ranging depth equal to one over the number of detectors(1/N). This scanning feature is different from the conventional delayscanning schemes used in the known LCI and OCT systems. The referencearm 303 optionally has a phase modulator mechanism, such as but notlimited to, an acoustooptic modulator, electro-optic phase modulator orthe like, for generating a carrier frequency.

Phase tracking is preferable performed to eliminate phase instabilitiesin the interferometer. Phase instabilities can cause individualinterferometric fringes to shift in location. If detection is slowrelative to the shifting of the fringes, the resulting averaging resultsin chirping of the interference signal. A-scan rate of 10 to 40 kHzresults in an effective integration time of 100 to 25 μs. Phaseinstabilities arising on a time frame shorter than the integration timeshould be compensated. Phase locking circuitry is commonly used inelectronics, and is frequently used in radar and ultrasound. Activephase tracking can be implemented by modulating the interferometer pathlength difference at 10 MHz with an electro-optic phase modulator in thereference arm over a fraction of the wavelength. By demodulating theintensity measured by one detector at the output of the interferometerat the frequency of the path length modulation, an error signal can begenerated indicating in which direction the phase modulator should shiftto lock onto a fringe amplitude maximum. By adding an offset to thephase modulator as determined by the error signal, the phase trackeractively locks onto a fringe maximum.

The phase modulator can modulate the path length difference over a fewwavelengths. The processing unit can determine if the phase modulatorhas reached its range limit, and jump by a full wave in phase tomaintain lock on a different fringe maximum. This approach exploits thefact that phase should be controlled only modulo 2π. In addition, theprocessing drives a slower component (e.g., the Rapid Scanning OpticalDelay (“RSOD”) line) to extend the path length range of the phasemodulator/RSOD combination over several millimeters. Phase locking canbe performed on a fringe maximum, minimum, or zero crossing, based onthe type of mixing performed in the demodulation circuit.

Another exemplary embodiment of the present invention can also useautoranging techniques and technology, including processing techniquesdescribed in U.S. patent application publication no. 2002/0198457, thedisclosure of which is hereby incorporated herein by reference in itsentirety. The autoranging mechanism may, in one exemplary embodiment,include a processor unit for (a) obtaining a first scan line; (b)locating a surface location “S” of a sample; (c) locating an optimalscan range “R” of the sample; (d) modifying a reference arm delaywaveform to provide an output; (e) outputting the output to a referencearm; (f) determining whether the image is complete; and/or (g) moving tothe next scan line if the image is not complete or remapping the imageusing the surface S data and the waveform data stored in the memorystorage device if the image is complete.

If the light signal returned from the sample has a low amplitude, phaselocking may be unstable due to the presence of noise. In anotherexemplary embodiment of the present invention, a separate, preferablymonochromatic, light source can be transmitted into the interferometer.The separate source wavelength may be within the wavelength tuning rangeof the OFDI source or may be centered at a different wavelength than theOFDI source spectrum. The separate source is preferably of higher power,and may be combined with the source arm (using wavelength division,multiplexer, grating, prism, filter or the like) travel to the referenceand sample arms and return back to the beam recombining element. Thereturned separate source light can then be separated from the OFDI lightfollowing transmission back through the beam recombining element (i.e.beam splitter output). A separation arrangement can perform spectralseparation by a dispersing element, such as a dichroic mirror, filter,grating, prism, wavelength division multiplexer or the like. Theseparate source will be detected separately from the OFDI light usingone or more detectors.

The higher power provided by this separate source can enable detectionof a higher amplitude interference pattern, and provide an improvedinput to the phase tracker, thus enabling more stable phase tracking

Referring now to FIG. 11, an in vivo image of a subject's fingertip(300*500 pixels) acquired at an A-line scan rate of 15.7 kHz is shownusing the exemplary embodiment of the system and process according tothe present invention. The optical sensitivity was measured to be about−100 dB. The SNR is superior to an equivalent TD OCT of the same A-linescan rate. The vertical line noise arises due to an un-optimizeddetection when there is a strong mirror-like reflection from the surfaceof the tissue, but should preferably be eliminated substantially by adetection optimization and/or an appropriate signal processing.

FIG. 12 shows another exemplary embodiment of a phase tracker system 600according to the present invention having an extended phase lock rangeis provided. This done by combining a fast element 602 (which may beprovided, for example, as an electro-optic (EO) phase modulator 602) tomodulate the path length difference over a small range, and a slowerelement 604 (which may, for example, be provided as a Rapid ScanningOptical Delay (RSOD) line 604) to modulate the path length over anextended range. The detector 606 signal can be mixed with the phasemodulator modulation frequency 608 by a mixer 610 and low pass filtered(filter not shown) to generate an error signal. The processing unit 612preferably processes the error signal to generate an offset voltage, andadds this offset voltage to the modulation signal 608, so as to generatethe output for the phase modulator driver 614. In addition, theprocessing unit 612 can generate a signal to the RSOD 604 to provideextended range tracking of the phase over distances of severalmillimeters. Light source 616, fiber splitter 618, sample arm 620 andreference arm 622 are shown, and are described herein.

The intensity I(t) at the detector at a given moment within a singleoscillation of the fringe pattern is given by

I(t)=cos[φ(t)]

where the phase φ gives the position in the fringe. For φ=0, the signalis at a fringe maximum, for φ=π, the signal is at a fringe minimum. Atan arbitrary moment t, the phase φ(t) is given by,

φ(t)=α+βsin(ωt)

where α describes the position within a single oscillation of the fringepattern, and β*sin(ωt) is the phase modulation introduced by the phasemodulator, with β the amplitude of the phase modulation, and ω thefrequency of the phase modulation signal. The intensity at thephotodetector I(t) can be mixed with a carrier at frequency ω and 2ω,resulting in the mixer signal MixerC(t), MixerS(t), Mixer2ωC(t) andMixer2ωS(t),

MixerC(t)=cos(ωt)*cos(α+βsin(ωt));

MixerS(t)=sin(ωt)*cos(α+βsin(ωt));

Mixer2ωC(t)=cos(2ωt)*cos(α+βsin(ωt)); Mixer2ωS(t)=sin(2ωt)*cos(α+βsin(ωt)).

The time average over a single oscillation of the carrier frequency w ofMixerC, MixerS, Mixer2ωC and Mixer2ωS is given by, MixerC(t)=0;MixerS(t)=sin(α)*J₁(β); Mixer2ωC(t)=cos(α)*J₂(β); Mixer2ωS(t)=0, whereJ₁(β) and J₂(β) are a Bessel functions of the first kind; its valuedepends on β, the amplitude of the phase modulation. Thus, the signalMixerS(t) and Mixer2ωC(t) are proportional to sin(α) and cos(α),respectively, with a the position within a single oscillation of thefringe pattern. The mixer outputs MixerS(t) and Mixer2ωC(t) are used asan error signal to generate an offset voltage to steer the phasemodulator to a new center position that minimizes the error signal, andlocks the interferometer output on a fringe maximum or minimum, or azero crossing, respectively. The complex spectral density can now bedetermined by two consecutive tuning scans, one where the error signalsin(α) is minimized, and the next where the error signal cos(α) isminimized, resulting in a 90 degrees phase shift between the twointerference patterns. Using this mixing arrangement, the complexspectral density can be obtained rapidly and without resorting to anadditional mechanical arrangement for changing the phase of thereference arm light.

FIG. 13 shows a further exemplary embodiment of an OFDI system 700 whichincludes a phase tracker for providing balanced detection according tothe present invention. In this exemplary embodiment, a source 702provides an electro-magnetic radiation (e.g., light) which passesthrough a splitter 704, that sends part of the light to a sample probe706 and the remainder of the light to a Rapid Scanning Optical Delay(“RSOD”) line 708. Light is passed from the RSOD 708 to the phasemodulator PM 710. Light from the phase modulator PM 710 is transmittedthrough a splitter 712, and then through two additional splitters 714and 716, a portion of the output of which is sent as balanced detectionoutputs to spectral detection units (not shown, but as describedelsewhere herein) and the remainder of the output is sent to the phasetracker assembly 720. In the phase tracker assembly 720, phase trackerdetectors D₁ and D₂, 722 and 724, receive the partial output of the pairof splitters 714 and 716, which in turn send signal to a mixer 726 togenerate an error signal. A processing unit 728 processes the errorsignal, where the sum generation of offset voltage and adds this to themodulation signal 730 to generate the output for the phase modulatordriver 732. Modulation signal, shown at box 730, is forwarded to themixer 726 and the processing unit 726. In addition, the fringe amplitudecould be too small for the phase tracker to lock. Alternatively, asecondary source with longer coherence length can be coupled to thesystem 700, e.g., to provide a larger fringe amplitude to the phasetracker.

FIGS. 14A-14C show an exemplary embodiment of a method for trackingphase in an imaging system begins in processing blocks 750 and 752according to the present disclosure invention by measuring a signalreceived from the sample arm (also see block 763 of FIG. 14B and block770 of FIG. 14C), and then increasing a phase of the signal (also seeblock 764 of FIG. 14B where the phase is adjusted and block 772 of FIG.14C where the phase is increased). Processing of this exemplary methodthen proceeds to block 754 of FIG. 14A, in which a first signalpartition of the signal defined as x₁ is measured at least one peak ofthe signal. In decision block 756, a determination is as to whether thesignal defined as x₁ has been measured at least one peak of the signal.If in decision block 756, it is determined that the signal defined as x₁has been measured at at least one peak of the signal, then processingreturns to block 754 and the signal is again measured.

On the other hand, if in decision block 756, it is determined that thesignal defined as x₁ has not been measured at at least one peak of thesignal, then processing flows to a decision block 758, where adetermination is made as to whether to adjust the signal. The adjustmentmay be, e.g., an increase or a decrease in the phase of the signal by anincremental amount as shown in blocks 760 and 762. Regardless of whetheran increase or a decrease in the phase of the signal is made, processingreturns to processing block 754, where a second signal partition of thesignal is measured at its peak. Blocks 756-762 are then repeated forsuch measured signal. It should be noted that the functions of blocks750-762 may be performed in parallel and/or series with other imagingprocesses.

The adjustment of phase “φ” can be defined as A(x₂-x₁), where “A” is aconstant and that the process of determining whether to increase ordecrease the phase of the signal by an incremental amount may furthercomprise the substeps of (1) determining whether A(x₂-x₁) is withinrange of the phase modulator; and (2) changing φ by an amount equal toA(x₂-x₁) if A(x₂-x₁) is within the range or changing φ by an amountequal to A(x₂-x₁)-m2π if A(x₂-x₁) is outside of the range, where M is aninteger greater than 1. The method may optionally further comprise asubstep (3) re-measuring signal x₁.

d. Data Processing

In general, the data recorded by the detector in time may not be sampledas a strictly linear function of the optical frequency w or wave numberk. The Fourier transform, however, can link z and k space (or t and w).Because of the non-linear sampling in k, the acquired spectrum isinterpolated to create evenly spaced samples in the k domain.Alternatively, the tuning slope of the laser could be adjusted in such away that the light is samples in equal intervals in k space, such thatthe interpolation becomes obsolete. Alternatively, the detection timingcould be designed to sample the light evenly spread in the k domain,such that the interpolation becomes obsolete. To achieve the optimalpoint spread function, dispersion in the sample and reference arm of theinterferometer should preferably be balanced. Dispersion imbalance canalso be corrected by digital processing. Phase chirping induced bymotions can also be corrected by digital processing. For the motionartifact correction, the axial movement of the sample is measured, and aproper nonlinear mapping can be calculated from the velocity of themovement.

Various interpolation techniques are known to those having ordinaryskill in the art. This includes, but is not limited to, simple two-pointinterpolation, FFT zero-padding followed by two-point interpolation, andrigorous interpolation with the sinc function dictated by the Nyquisttheorem.

An exemplary embodiment of the present invention may also provide aprobe for locating atherosclerotic plaque in a blood vessel, comprising:an interferometer; a spectral separating unit which splits signalreceived from the interferometer into a plurality of opticalfrequencies; and a detector arrangement capable of detecting at least aportion of the optical frequencies received from the spectral separatingunit.

e. Frequency Shifting Technique

For high-speed OFDI techniques, the maximum ranging depth can likely belimited by the finite width of the coherence function of the laseroutput because the coherence length is often compromised to obtainhigher tuning speed, higher output power, or wider tuning range. Thefinite coherence length may cause the visibility of the interferencefringe to decrease as the path length difference of the interferometerincreases. This result in the degradation of SNR, and therefore limitsthe maximum ranging depth. Furthermore, the inability to distinguishbetween a positive and negative electrical frequency in a conventionalinterferometry may lead to the ambiguity between positive and negativedepths. To avoid the imaging folding artifact, the reference delay ofthe interferometer should be adjusted so that the image presents at onlyeither positive or negative depth. This further may limit the rangingdepth for a given coherence length of the source.

To avoid such possible limitation, quadrature interference signals havebeen measured based on active or passive phase biasing using apiezoelectric actuator, birefringence plate or 3'3 coupler. Thesetechniques may provide otherwise overlapping images associated withpositive and negative depths, but tended to leave significant residualartifacts due to the difficulty of producing stable quadrature signals.In this paper, we propose and demonstrate a simple technique thateffectively eliminates the ambiguity between positive and negativedepths.

The exemplary technique according to the exemplary embodiment of thepresent invention uses an optical frequency shifter in theinterferometer to provide a constant frequency shift of the detectorsignal. This allows both sides of the coherence range to be used withoutcrosstalk, and can double the ranging depth. The same concept has beendescribed above in the context of 1-dimensional optical frequency domainreflectometry using rotating birefringence plates at 58 Hz or arecirculating frequency shifting loop. In this exemplary embodiment, anacousto-optic frequency shifter is used, and the technique is applied tohigh-speed OFDI with several orders of magnitude faster ranging speed.Furthermore, a signal processing technique according to a furtherexemplary embodiment of the present invention is provided to accommodatea nonlinear tuning slope of the swept source in the frequency shiftingtechnique.

A. Principle Frequency Shift

FIG. 15 shows a high level diagram of the OFDI system according to thepresent invention which includes a wavelength-swept source 95,singlemode-fiber interferometer employing an optical frequency shifter311 in a reference arm 80, a photodetector 88, and a signal processor160. With a roundtrip frequency shift of Δf in the reference arm, thephotocurrent associated with the interference between the reference andsample light can be expressed as

${{i_{s}(t)} = {\eta \sqrt{{P_{r}(t)}{P_{s}(t)}}{\int{\sqrt{R(z)}{G\left( {z} \right)}{\cos \left\lbrack {{\frac{4\pi}{c}{v(t)}z} + {\varphi (z)} + {2{\pi\Delta}\; {ft}}} \right\rbrack}{z}}}}},$

where η denotes the quantum efficiency of the detector, Pr(t) and Ps(t)the optical powers of the reference and sample arm light, respectively,R(z) the reflectivity profile of the sample, G(|z|) the coherencefunction corresponding to the fringe visibility, c the speed of light,ν(t) the optical frequency, and φ(z) the phase of backscattering. In thecase of a linear tuning, i.e. v(t)=0-1 t, the frequency of the detectorsignal is given by

$f_{s} = {{{v_{1}\frac{2\; z}{c}} - {\Delta \; f}}}$

The zero signal frequency corresponds to a depth z=cΔf/(2 ν1).Therefore, by choosing the direction of frequency shifting same as thetuning direction of the swept source, the zero signal frequency can bemade to point to a negative depth. FIGS. 16( a) and 16(b) illustrate theeffect of the frequency shift. The fringe visibility or the coherencefunction has a peak value at the zero depth and decrease as the depthincreases. The coherence length z_(c) indicates the depth where thevisibility drops to 0.5 and thereby the SNR drops by 6 dB. One mayarguably define the effective ranging depth as the maximum depth spanwhere the SNR penalty is less than 6 dB. For example, in FIG. 16( a), asingle side of the coherence range can be used due to the sign ambiguityof the signal frequency (hatched region). In contrast, as shown in FIG.16( b), with an appropriate frequency shift, both sides of the coherencerange from −z_(c) to z_(c) can be utilized without any image crosstalkbetween the negative and positive depths.

Nonlinear Tuning

Nonlinearity in ν(t) with respect to time results in frequency chirpingof the signal at a constant depth and causes the degradation of axialresolution. As a solution to this problem, the detector signal may besampled with nonlinear time interval compensating for the frequencychirping. Alternatively, the detector signal can be sampled with aconstant time interval, and then the sampled data be mapped to a uniformv-space by interpolation prior to discrete Fourier transform (“DFT”).Both methods have been demonstrated to yield a transform-limited axialresolution. However, these methods are not applicable directly in thefrequency shifting technique. Both the nonlinear sampling andinterpolation method can result in artificial chirping of the frequencyshift, leading to sub optimal axial resolution. Thus, a modifiedinterpolation method can be used to achieve nearly transform-limitedaxial resolution over the entire ranging depth. The exemplary techniquemay be as follows:

-   Step 1. Obtain N samples of the signal with uniform time interval    during each wavelength sweep of the source.-   Step 2. Produce DFT of N data points in the electrical frequency    domain.-   Step 3. Separate two frequency bands below and above Δf    corresponding to negative and positive depths, respectively.-   Step 4. Shift each frequency band such that the zero depth is    aligned to the zero electrical frequency.-   Step 5. Apply zero-padding to each frequency band and perform    inverse DFT resulting in an array of increased number of samples in    the time domain with smaller time interval for each frequency band.-   Step 6. Interpolate each array in the time domain into a uniform v    space using a proper mapping function given by the nonlinearity of    the swept source.-   Step 7. Conduct DFT of each interpolated array.-   Step 8. Combine the two arrays (images) by shifting the array index.

As a result, the zero depth lies at the electrical frequency of Δf.

B. Experiment OFDI System

FIG. 17 depicts the experimental setup of an exemplary OFDI systememploying two acousto-optic frequency shifters (FS1 800 and FS2 802,Brimrose Inc. AMF-25-1.3) according to an exemplary embodiment of thepresent invention. The two frequency shifters may be driven with voltagecontrolled oscillators to produce a net shift of Δf=FS2-FS1. The use oftwo frequency shifters balanced the material dispersion of theacousto-optic crystals automatically. The insertion loss of each deviceincluding fiber coupling may be less than 2.5 dB. The sampling rate ofthe digitizer can be 10 MHz. The swept laser 100 may be constructed toprovide a tuning range of 108 nm centered swept from 1271 nm to 1379 nm(ν1=135 GHz/μs). Although a repetition rate up to 36 kHz could beachieved, the laser was operated at a reduced rate of 7 kHz and 1300samples were acquired during a single wavelength sweep. This resulted ina depth span of 5.8 mm in the image corresponding to the Nyquistfrequency of 5 MHz. The probe 810 may include a galvanometer mirror andan imaging lens produced a probe beam with a confocal parameter of 1.1mm. An optical tap coupler 820 can be used in conjunction with anarrowband filter 830 and a photodetector 834 to generate a TTL triggersignal in an electrical circuit 836. The TTL signal may be used as atrigger in analog to digital conversion.

The interference signal can be measured using a dual balanced receiver151. The detector signal was further processed prior to digitizationusing a low pass electrical filter 840. Other types of electricalfilters such as a band pass filter and a slope filter. The transmissionof the slope filter may have an exponentially-rising (falling) slope inits low frequency band. This filter may be useful in which attenuationis significant and the resulting signal strength decays with depth. Theslope filter can improve the dynamic range of the detection byeffectively suppressing the large signal from the surface relative tothat at greater depths.

To characterize the coherence function of the swept laser 100, the pointspread function of the system may be measured at Δf=0 (FS1=−25 MHz,FS2=−25 MHz) with a partial reflector at various locations of thereference mirror. For comparison, the sampled data acquired at eachdepth was processed with and without the mapping process. FIGS. 18( a)and 18(b) show exemplary results, where the y-axis represents the squareof the DFT amplitudes normalized to the value at zero frequency, and thebottom and top x-axes represent the signal frequency and the depth z,respectively. Without mapping, the point spread function suffers fromsignificant broadening and large degradation of the peak power as thedepth increases, because of the nonlinearity of our swept laser [seeFIG. 18( a)]. With the mapping process, however, the spread functionexhibits nearly transform-limited axial resolution as shown in FIG. 18(b). The finite coherence length of the laser output accounts for thedecrease of the signal power depth. Over the entire depth span of 5.8mm, the SNR is reduced by more than 11 dB. According to the criterionfor the effective ranging depth introduced earlier, the depthcorresponding to the coherence length may be only 2.9 mm, a half thetotal in the image. The same experiment was conducted with a nonzerofrequency shift of Δf=−2.5 MHz (FS1=−22.5 MHz, FS2=−25 MHz). FIGS. 18(c) and 18(d) show the point spread functions measured with and withoutthe mapping process, respectively. As shown in these figures, the peakof the signal power occurring at the zero depth present at a frequencyof 2.5 MHz is at least approximately equal to the net acousto-opticfrequency shift. The nearly transform-limited axial resolution observedin FIG. 18( d) validates the mapping technique. The reduction in thesignal power is less than 5 dB over the entire depth span of 5.8 mm,demonstrating the benefit of the frequency shifting technique in termsof extending the ranging depth.

Image

Exemplary imaging of a human lung tissue ex vivo was conducted with theOFDI system. FIGS. 19A and 19B depict two images/graphs, obtained underidentical experimental conditions except that Δf=0 for the image/graphin FIG. 19A and Δf=−2.5 MHz for the image/graph in FIG. 19B. Eachimage/graph was obtained using the mapping technique described above.The surface of the tissue was placed with an angle with respect to theprobe beam axis, and the reference mirror was positioned such that thesignal was present at both positive and negative depths in the image. InFIG. 19A, the tissue image is contained within the effective rangingdepth of 2.8 mm, i.e. the top half of the total depth span. However, therelatively large variation in the sample location resulted in theimaging folding artifact. In contrast, in FIG. 19B the entire positiveand negative depths could be displayed without ambiguity takingadvantage of the ranging depth increased to 5.8 mm by the frequencyshifting technique.

The foregoing merely illustrates the principles of the invention.Various modifications and alterations to the described embodiments willbe apparent to those skilled in the art in view of the teachings herein.It will thus be appreciated that those skilled in the art will be ableto devise numerous systems, arrangements and methods which, although notexplicitly shown or described herein, embody the principles of theinvention and are thus within the spirit and scope of the presentinvention.

1-93. (canceled)
 94. An apparatus, comprising: at least oneradiation-providing structural first structural arrangement configuredto provide a radiation which includes at least one firstelectro-magnetic radiation directed to a sample and at least one secondelectro-magnetic radiation directed to a reference, wherein a frequencyof the radiation provided by the at least one first arrangement variesover time; and at least one detector second arrangement configured todetect an interference between at least one third radiation associatedwith the at least one first radiation and at least one fourth radiationassociated with the at least one second radiation to generate anelectrical first signal, wherein the at least one second arrangement isconfigured to obtain a second signal associated with at least one phaseof at least one frequency component of the electrical signal, andwherein the at least one second arrangement includes a computingarrangement which compares the second signal to at least one particularinformation.
 95. The apparatus according to claim 1, wherein the atleast one second arrangement is configured to determine a third signalassociated with at least one further phase of at least one furtherfrequency component of the electrical signal, and wherein the at leastone particular information is the third signal.
 96. The apparatusaccording to claim 1, wherein the at least one second arrangement isconfigured to determine a third signal associated with at least onefurther phase of at least one further frequency component of a furtherelectrical signal, the further electrical signal being different fromthe electrical signal, and wherein the at least one particularinformation is the third signal.
 97. The apparatus according to claim96, wherein the electrical signal and the further electrical signal areobtained at different times.
 98. The apparatus according to claim 96,wherein the electrical signal and the further electrical signal areobtained at different locations of the sample.
 99. The apparatusaccording to claim 94, further comprising at least one structural thirdarrangement is configured to generate a third signal associated with theradiation, wherein the at least one second arrangement is configured toprovide the particular information based on the third signal.
 100. Theapparatus according to claim 99, wherein at least one third arrangementincludes an interferometric arrangement.
 101. The apparatus according toclaim 94, further comprising at least one structural third arrangementconfigured to receive at least one portion of the radiation, and providea further radiation so as to generate a third signal, wherein at leastone of the second signal or the particular information is associatedwith the further radiation.
 102. The apparatus according to claim 94,wherein the at least one second arrangement configured to detect a thirdsignal corresponding to the electrical signal between the at least onesecond and third radiations, and wherein the second signal is associatedwith at least one phase of at least one frequency component of anadditional electrical signal associated with the third signal.
 103. Amethod comprising: providing a radiation which includes at least onefirst electro-magnetic radiation directed to a sample and at least onesecond electro-magnetic radiation directed to a reference, wherein afrequency of the radiation varies over time; detecting an interferencebetween at least one third radiation associated with the at least onefirst radiation and at least one fourth radiation associated with the atleast one second radiation to generate an electrical first signal;obtaining a second signal associated with at least one phase of at leastone frequency component of the electrical signal; and comparing thesecond signal to at least one particular information.
 104. The methodaccording to claim 103, further comprising determining a third signalassociated with at least one further phase of at least one furtherfrequency component of the electrical signal, and wherein the at leastone particular information is the third signal.
 105. The methodaccording to claim 104, further comprising: determining a third signalassociated with at least one further phase of at least one furtherfrequency component of a further electrical signal, the furtherelectrical signal being different from the electrical signal, whereinthe at least one particular information is the third signal.
 106. Themethod according to claim 105, wherein the electrical signal and thefurther electrical signal are obtained at different times.
 107. Themethod according to claim 105, wherein the electrical signal and thefurther electrical signal are obtained at different locations of thesample.
 108. The method according to claim 104, further comprisinggenerating a third signal associated with the radiation; and providingthe particular information based on the third signal.
 109. The methodaccording to claim 103, wherein the second signal is an interferometricsignal.
 110. The method according to claim 103, further comprising:receiving at least one portion of the radiation and provide a furtherradiation, wherein at least one of the second signal or the particularinformation is associated with the further radiation.
 111. The methodaccording to claim 103, further comprising detecting a third signalcorresponding to the electrical signal between the at least one secondand third radiations, wherein the third signal is associated with atleast one phase of at least one frequency component of an additionalelectrical signal associated with the third signal.
 112. A systemcomprising: at least one arrangement configured to (i) detect aninterference between at least one first radiation associated with afirst split portion of a radiation received from a sample and a secondsplit portion of the radiation received from a reference and (ii) togenerate an electrical first signal associated with the interference,wherein a frequency of the radiation varies over time, and wherein theat least one arrangement is configured to obtain a second signalassociated with at least one phase of at least one frequency componentof the electrical signal, and wherein the at least one arrangementincludes a computing arrangement which compares the second signal to atleast one particular information.
 113. The apparatus according to claim112, wherein the at least one first arrangement is configured to detecta third signal corresponding to the electrical signal between the firstand second split portions of the radiation, and wherein the secondsignal is associated with at least one phase of at least one frequencycomponent of an additional electrical signal associated with the thirdsignal.